Snubber circuit

ABSTRACT

A snubber circuit comprises a first energy storage device and circuitry coupled to the first energy storage device to facilitate capturing, by the first energy storage device, energy of a switching circuit. The snubber circuit also comprises a second energy storage device coupled to the first energy storage device to store the captured energy. The circuitry additionally facilitates resetting of the first energy storage device.

CROSS-REFERENCE TO RELATED APPLICATION

This application claims priority from copending applications having Ser.No. 10/763,664, entitled “ALTERNATING CURRENT SWITCHING CIRCUIT”) andSer. No. 10/764,409, entitled “POWER CONVERTER”) each of which werefiled on Jan. 23, 2004 and each of which are hereby incorporated byreference herein.

BACKGROUND

Alternating Current (AC) circuits comprising inductive loading containstored energy that, when the circuit is switched off, needs to bedissipated. If this stored energy is not accounted for in the design ofthe circuit, the result could be a number of undesired effects on thecircuit and/or the circuit's surrounding environment.

One undesired effect on the circuit can be the build-up of heat in acircuit. For example, circuitry utilized in a switching device may heatup. This may result in requiring a designer to include a heat sink for aswitching device. The addition of a heat sink may add cost to a design.

Another undesirable effect on a circuit with stored inductive energy isthat the switching-off of the circuit could result in large dischargetransients being dissipated throughout the rest of the circuit. Theselarge discharge transients may cause damage to other circuit elementsthat absorb the energy of the discharge transients.

Yet, another undesired effect may be radio frequency (RF) emissions overa desired level. Various jurisdictions classify devices and limit thetypes of devices that can be sold. For example, in the United States,the FCC certifies devices as “Class A” or “Class B” depending on theamount of RF energy that the device emits. “Class B” devices areauthorized for home use whereas “Class A” devices are limited to officeuse.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments of the present invention will be described by way ofexemplary embodiments, but not limitations, illustrated in theaccompanying drawings in which like references denote similar elements,and in which:

FIG. 1 illustrates an AC MOSFET switch, including anti-parallel diodes,in accordance with one embodiment.

FIG. 2 illustrates a more detailed look at an AC MOSFET switch,including intrinsic parasitic diodes of the MOSFETs, in accordance withone embodiment.

FIG. 3 illustrates current that is delivered to a load when oneembodiment of the AC MOSFET switch is utilized to control current.

FIGS. 4A-4C illustrate a power filter and its effects on the currentdrawn by a load driven by an AC MOSFET switch, in accordance with oneembodiment.

FIG. 5 illustrates an AC MOSFET switch design including a snubbingdevice, in accordance with one embodiment.

FIG. 6 illustrates a single IC device containing two NMOS type MOSFETdevices of an AC MOSFET switch, in accordance with one embodiment.

FIG. 7 illustrates an imaging device, suitable for housing an apparatusutilizing a snubber circuit, in accordance with one embodiment.

FIG. 8 illustrates a fuser power control circuit utilizing an AC MOSFETswitch including a regenerative snubber, in accordance with oneembodiment.

FIG. 9 illustrates combined snubber and bias circuitry, in accordancewith one embodiment.

FIG. 10 illustrates combined snubber and bias circuitry, in accordancewith another embodiment.

FIG. 11 illustrates a combined snubber and bias circuit, in accordancewith yet another embodiment.

FIG. 12 illustrates a regenerative snubber, in accordance with anotherembodiment.

FIG. 13 illustrates a regenerative snubber with additional DC bias, inaccordance with another embodiment.

FIG. 14 illustrates a regenerative snubber in use with a DC-DCconverter, in accordance with one embodiment.

DETAILED DESCRIPTION OF THE EMBODIMENTS

Although specific embodiments will be illustrated and described herein,it will be appreciated by those of ordinary skill in the art that a widevariety of alternate and/or equivalent implementations may besubstituted for the specific embodiments shown and described withoutdeparting from the scope of the present invention. This application isintended to cover any adaptations or variations of the embodimentsdiscussed herein. Therefore, it is manifestly intended that thisinvention be limited only by the claims.

The following discussion is presented in the context of MOSFET devices.It is understood that the principles described herein may apply to othertransistor devices.

Refer now to FIG. 1 wherein an AC MOSFET switch 110, includinganti-parallel diodes 112 114, is illustrated, in accordance with oneembodiment. For the MOSFETs 142 144 illustrated, the sources of theMOSFET devices are coupled at junction 102. In one embodiment, MOSFETs142 144 are power MOSFETs. In addition, the gates are electricallycoupled at junction 104. These couplings are to facilitate the operationof the two MOSFETs 142 144 as a single AC MOSFET switch. Thus, byapplying a gate to source voltage, V_(GS), greater than the thresholdvoltage, V_(TH), to the two MOSFETs 142 144, both MOSFETs conductcurrent 120.

Also illustrated in FIG. 1 are two diodes 112 114. These diodes 112 114,which may be parasitic or explicit, are anti-parallel to theirrespective MOSFETs. As described in further detail below, these diodes112 114 may be utilized to bypass the intrinsic anti-parallel diodes ofthe MOSFETs. Thus, as illustrated, the anodes of the diodes 112 114 arecoupled to the sources of the diodes' respective MOSFET and the cathodesare coupled to the respective drains.

FIG. 1 also illustrates the AC MOSFET switch in use in controlling powerto a load. As previously mentioned, AC MOSFET switch 110 comprises twoMOSFETs 142 144. AC MOSFET switch 110 controls current 120 through load130. This may be accomplished by switch control circuit 140 whichapplies the gate-source voltages for the two MOSFETs 142 144 forming theAC MOSFET switch 110. In the embodiment illustrated, charge pump biasingcircuit 150 supplies current to switch control circuit 140 from line (L)172 and neutral (N) 174 connections of the AC power source.

FIG. 2 illustrates a more detailed look at an AC MOSFET switch,utilizing P type MOSFETs, including intrinsic parasitic diodes 232 234of the MOSFETs 242 244, in accordance with one embodiment. Alsoillustrated are antiparallel diodes 212 214 which may be utilized tobypass the intrinsic anti-parallel diodes 232 234 of the MOSFETs. Notethat the sources of both MOSFETs 242 244 are coupled 204 to each other.In addition, the gates of both MOSFETs 242 244 are coupled 206 to eachother. When a voltage, V_(SG) 280 less than a threshold voltage V_(TH)is applied, the MOSFETs 242 244 will be “turned-off” and the internalreverse biased PN junctions will substantially prevent current fromflowing through the MOSFETs.

When a voltage, V_(SG) 280 greater than a threshold voltage V_(TH) isapplied to the common sources and gates of MOSFETs 242 244 are turned onto facilitate the flow of current through the AC MOSFET switch. Notethat current will flow in the reverse direction in MOSFET 242 or 244depending on the polarity of the AC voltage source. That is, in thereverse direction as is normally used in DC circuits, that is drain tosource in an N type MOSFET or source to drain in a P type MOSFET. Thereverse current flow causes no problem as the MOSFET transistor is trulya bidirectional device, that is, current may flow from drain to sourceor source to drain once the proper gate voltage is applied and theconductive channel forms. Normally, during reverse polarity across thesource/drain of a MOSFET, an internal PN junction, represented byparasitic diodes 234 and 232 in FIG. 2, will eventually turn on allowingcurrent 271 to flow. Note that parasitic diodes 234 and 232 are notseparate from the MOSFET 244 and 242; e.g. parasitic diode 234 is a PNjunction that is part of the structure of transistor 244. Once the gatevoltage is removed the parasitic diode conducts during reverse currentflow which makes a single MOSFET unsuitable for the control ofalternating current 271 273. The common source configuration of MOSFET242 and 244 of FIG. 2 results in one of the parasitic diodes in areverse biased state which substantially prevents current flow throughthe parasitic diodes 232 234 when the MOSFETs are in either theconducting or nonconducting states.

Referring again to FIG. 1, switch control circuit 140 and charge pumpcircuitry 150 are utilized to provide control for the application of thevoltage to the gates of MOSFETs 142 144. In the embodiment illustrated,switch control circuit 140 may be an externally controlled pulse widthmodulation circuit. In the embodiment illustrated, charge pump 150utilizes the AC line to power the pulse width modulation circuitry. Inaddition, the frequency of the modulated control signal may be fixed,whereas the duty cycle of the modulation, as described below, isutilized to determine the power to be delivered to the load 130. In analternative embodiment the gate and source of the AC MOSFET may bedriven by a circuit which has a minimum conduction time combined with avariable frequency to determine the power to be delivered to the load130.

FIG. 3 illustrates current that is delivered to a load when oneembodiment of the AC MOSFET switch is utilized to control current. Forexample, as discussed above with respect to FIG. 1, the switch controlcircuit 140 may be a pulse width modulation circuit. In such a case, thepower delivered to the load 130 can be controlled by changing the dutycycle of the pulse control signal. FIG. 3 illustrates an example inputvoltage 310 from the line and neutral. Illustrated also, in the darkshaded regions 320, are the periods where the AC MOSFET switch 110 isswitched on to allow current to flow through the load 130. The voltage310 and current 320 are normalized so that they share a common envelope.Thus, in the illustrated embodiment, a 50% duty cycle signal driving thegate to source voltage will result in an effective power of one half thetotal power available being delivered the load. By utilizing a pulsewidth modulation technique, the level of power delivered to the load canbe adjusted by controlling the width of the pulses generated by thepulse width modulation of the switch control circuit. The equationgoverning the power transfer to the load is:

${Pavg} = {\frac{{Vrms}^{2}}{R} \cdot {d.}}$Where V_(rms) is the Root Mean Square (rms) voltage of the AC powersource, R is the resistance of the load and d is the duty ratio of thepulse width modulator driving the AC MOSFET. By inspection of thisequation, the power transferred to the load is a linear function of theduty ratio of the pulse width modulator. The load is at zero power whenthe duty ratio is zero and at maximum power when the duty ratio is 1.

In an alternative embodiment in which the gate and source of the ACMOSFET switch are driven by a circuit which has a minimum conductiontime combined with a Variable Frequency Oscillator (VFO) the powerdelivered to the load 130 is determined byP=V ² ÷R×f×T _(min)Where V is the rms voltage of the AC power source, R is the resistanceof the load, f the frequency of the VFO driving the AC MOSFET andT_(min) the minimum conduction time allowed. By inspection, thisequation shows that the power transferred to the load is a linearfunction of the frequency of the VFO. The load is at zero power when theVFO frequency is 0 and at maximum power when the period of the frequencyof the VFO is equal to or less than the minimum allowed conduction timeT_(min).

The above examples operate to facilitate the switching of thealternating current at relatively higher frequencies. There areadvantages to switching the current at relatively higher frequencies.Switching frequencies out of the audio range (e.g. greater than 20 KHz)can be utilized to reduce human factor issues associated with audibleswitching noise. Another advantage of operation at higher frequenciesmay be a reduction in switching and conduction losses. Implementationsoperating at significantly lower frequencies spend more time in thelinear region of operation. Spending more time in the linear regionduring switching may dissipate significant amounts of additional energyin the form of heat as relatively slow transitions are made through thislinear region. In addition, because of the relatively low voltage dropsassociated with the disclosed switching of alternating current, lessenergy is dissipated from the product of the current flowing across thevoltage drops of the devices. In addition, the AC MOSFET switchingcircuit above does not introduce significant harmonics into thealternating current. This can reduce costs associated with filteringthese harmonics to meet international regulatory requirements.

FIG. 4A illustrates input circuitry for an AC MOSFET switch, inaccordance with one embodiment. Illustrated is a filter stage 410 toprovides a high frequency short to ground to any transients or conductedemissions that occur across the inputs. Illustrated also is a filteringstage 420 to provide smoothing of the alternating current drawn by theload 430. The effect of this filter is to smooth the harmonic richcurrent drawn by the pulse width modulated, or VFO driven load, suchthat the power source experiences a continuous current flow withvirtually no harmonic current content.

In the embodiment, switch control circuit 450 switches the current 472delivered to the load as illustrated in FIG. 4B. During times ofswitching, assuming a purely resistive load, the current 472 through theload 430 will follow the line voltage provided, that is, it will be inphase. When the switch is turned off, the current delivered to the loadwill drop to zero 474. Thus, as can be seen there will be dramaticshifts or steps in the current drawn by the load as the switch turns onand off. These step changes in the current represent unwanted currentharmonics placed on the AC power source which may exceed regulatorylimits. To solve this problem, filtering stage 420 is added to thecircuit. FIG. 4C illustrates the current drawn from the AC power sourceat the line and neutral connections by the switched load as a result ofthe filtering stage 420. When the switch is turned off, the filteringstage 420 smoothes current 476 drawn by the load 430. In the case inwhich the switch is driven by a pulse width modulator, the totalinstantaneous current drawn by the circuit may be the sum of thefundamental current and the instantaneous value of the ripple current.This instantaneous current may be expressed as

${i_{L}(t)} = {{\frac{V \cdot d}{R} \cdot {\sin( {2 \cdot \pi \cdot f_{o} \cdot t} )}} + {\frac{\pi^{2}}{4} \cdot ( {1 - d} ) \cdot ( \frac{f_{c}}{f_{s}} )^{2} \cdot \frac{V \cdot d}{R} \cdot {\sin( {2 \cdot \pi \cdot f_{o} \cdot t} )} \cdot {{\sin( {2 \cdot \pi \cdot f_{s} \cdot t} )}.}}}$where f_(C) is the resonant frequency of filtering stage 420, f_(S) isthe switch frequency of the pulse width modulator, f_(O) is thefrequency of the AC power source, d is the duty cycle of the pulse widthmodulator, V is the peak source voltage, and R is the load resistance430. Under direct examination of this equation it is noted that, as theswitch frequency of the pulse width modulator is increased, theresultant alternating current waveform at the Line and Neutralconnections smoothes dramatically.

FIG. 5 illustrates an AC MOSFET switch design including a snubbingdevice 580, in accordance with one embodiment. Snubbing device 580 isutilized for dissipating energy stored in the circuit. Stored energy ina circuit exists due to various factors associated with the circuit suchas: parasitic inductance associated with the wiring providing the ACcurrent, parasitic inductance in the components leads, and inductance inthe load itself. Snubber designs are designed to capture a portion ofthe stored energy in a circuit, when the circuit is switched off. Thesesnubber designs are to reduce, among other things, the resonance of thecircuit. However, these snubber designs are not engineered to dissipateall the energy; they are simply designed to dissipate enough energy toreduce resonance and the resulting resonant “over” voltages that mayotherwise occur.

To dissipate all the energy in the circuit, a significantly larged sizedcapacitor 573 may be used in snubber 580 design. It is desirable to havethe resistance 577 approximately match the resistance in the load 530.Thus, if the load resistance is approximately 20 ohms, then theresistance of the snubber should be selected to be about 20 ohms. Inaddition, the stored inductance 575 for a typical circuit driving the ACMOSFET switch has been measured at approximately 100 nanoHenries. Insome snubber designs, a capacitor capable of capturing about ⅕ of theenergy stored in the inductive parasitics may be utilized. As mentioned,this capacitor size is utilized to simply avoid resonance of thecircuit. However, the remaining energy is dissipated via heat in theswitching element or as Radio Frequency (RF) emissions. To avoid thisheat or RF emissions, a larger snubber circuit may be utilized.

In order to have the snubber dissipate substantially all the storedenergy of the circuit, the energy dissipated by the snubber should equalthe energy stored due to the inductance of the circuit. Thus,½LI ²=½CV ²,where I=V/R½L(V/R)²=½CV ²Solving for C we find that:C=L/R ²Thus, the capacitor used is directly related to the value of theparasitic inductance.

Dissipating heat may be undesirable as it may result in damage to thecircuit. A solution to this may be to include a heat sink. However, theaddition of the heat sink may add cost to the design. In addition,generation of RF emissions may be undesirable as it may result in poorclassification during RF certification proceedings for the devicecontaining the AC MOSFET switch. To protect from RF emissions, a shieldfor the RF emissions may be provided. Again, however, the addition of ashield may add cost to the design.

Thus, in one embodiment, the capacitor that is part of the snubberillustrated in FIG. 5 is designed to capture substantially all of thestored energy in the circuit associated with the AC MOSFET switch. Inthis manner, the design of RF shield and the design of any heatdissipating devices may be reduced.

FIG. 6 illustrates a single integrated circuit (IC) device 600containing two NMOS type MOSFET devices of an AC MOSFET switch, inaccordance with one embodiment. In an alternative embodiment, two PMOStype MOSFET devices may be utilized in the construction of an AC MOSFETswitch. Recall that the two sources from the two MOSFETs are logicallycoupled to each other in the AC MOSFET switch. By fabricating the twoMOSFETs in a single package on an IC, the two MOSFETs may share a commonsource region 610 on the IC. In the embodiment illustrated in FIG. 6, acommon source region 610 is implanted into the die containing the ACMOSFET switch. The sharing of the common source region 610 may allow theuse of a single source lead emanating from the package containing thetwo MOSFETs of AC MOSFET switch. This, in turn, may result in decreasedconduction resistance due to the elimination of one source lead and thesource lead's associated wire bonding parasitics, such as ohmicresistance from the die to a package lead. For example, in oneembodiment, the elimination of one of the source leads may reduce theimpedance by 70 milliohms, corresponding to the impedance associatedwith one of the leads to the AC MOSFET switch.

70 milliohms may be a substantial portion of the overall resistanceassociated with the AC MOSFET switch. For example, assume an R_(DSON) of100 milliohms for each MOSFET in the AC MOSFET switch. Thus, with a 70milliohm resistance for each lead for the source and drain, the overallpath impedance across the source and drain is 240 milliohms. Twodiscrete series devices have an effective resistance through the ACMOSFET switch of 480 milliohms. Recall that the external source lead inthe AC MOSFET is used for the application of gate bias and as aconduction path for certain types of snubber applications during switchturn off. By design the external source connection 610 has very lowcurrent flow and does not introduce series resistance to the AC MOSFETswitch when the switch is conducting. This fact allows the conductionresistance of the AC MOSFET switch to be reduced by 140 milliohms, or areduction in effective resistance 30% by using a common source region onthe die of the AC MOSFET and the elimination of one lead. Since thepower dissipated is directly related to the resistance, this results ina 15% reduction in power loss, for the embodiment described. Fabricationof the AC MOSFET switch on a single die also allows one of the gateterminals of the discrete implementation to be eliminated. The result ofthe common source region and eliminated gate terminal is a four pindevice with two high current drain connections and two lower currentgate and source connections. One pin of the four pin device is coupledto each of the gates of the two MOSFETs. Another pin is coupled to thecommon source region, and each of the two remaining pins are coupled toa different one of the drains.

Thus, embodiments of an AC MOSFET switch design have been disclosed.This design generally allows for faster operation of the AC MOSFETswitch to, among other things, allow operation significantly above theaudio frequency spectrum (e.g. greater than 20 kHz). The AC MOSFETswitch operation generally utilizes higher frequencies which, in turn,allows the device to be used in a broad range of AC power control, thusreducing the use of rectification and the resulting induction ofharmonics to the power line. These advantages reduce the use ofexpensive filtering and allow for better operation in environmentscontaining persons such as the home or office environment. The designsmay also allow for single IC design of the AC MOSFET switch in manyapplications. This may reduce the number terminal thus reducing loss dueto lead resistance.

While various circuit elements are illustrated, it is understood bythose skilled in the art that equivalent circuit elements can beutilized without altering the spirit of the embodiment disclosed. Forexample, in the place of a single bias capacitor, multiple parallelcapacitors may be utilized to obtain a desired effective capacitance.The term “capacitor” as used herein (in the specification and in theclaims) includes its common meaning as understood by those of ordinaryskill in the art, i.e. an electronic device with the ability of storingcharge, as well as other devices or combination of devices configured toprovide the ability to store charges.

The bias circuitry utilized to drive control circuitry of the AC MOSFETswitch may be combined with the snubber circuitry. By combining the biascircuitry with the snubber circuitry, power that may otherwise be wastedin the snubber circuitry may be utilized to drive the control circuitry.

FIG. 7 illustrates an imaging system 700, suitable for housing anapparatus utilizing a snubber circuit, in accordance with oneembodiment. As illustrated, for the embodiment, imaging system 700includes processor/controller 702, memory 704, imaging engine 706 andcommunication interface 708 coupled to each other via bus 710. Imagingengine 706 comprises a fusing subsystem 720 for fusing toner to paper.In addition to fusing subsystem, imaging system may comprise otherinductive heating elements or induction motors. Imaging engine 706 issimilar to those found in many imaging systems, such as those availablefrom Hewlett Packard Corp. of Palo Alto, Calif. Fusing subsystem 720 isconnected to an alternating current power source through interface 730.Fusing subsystem 720 may utilize a snubber circuit as described by thepresent disclosure.

Processor 702, in combination with other portions of the imaging system700, can perform various control functions of the fusing subsystem 720.For example, in one embodiment, processor 702 controls power managementof the fusing subsystem 720 to intelligently power down the fusingsubsystem when the fuser is not in use. Otherwise, processor 702, memory704, imaging engine 706, comm. interfaces 708, and bus 710 represent abroad range of such elements.

FIG. 8 illustrates a fuser power control circuit utilizing an AC MOSFETswitch 840 including a regenerative snubber 810, in accordance with oneembodiment. A control circuit 820, such as a linear analog pulse widthmodulator (PWM), controls power delivered to a fusing heating element830 by an AC MOSFET switch 840. As the control circuit 820 turns off theAC MOSFET switch 840, current is diverted through regenerative snubber810. Regenerative snubber 810 contains circuitry to generate biasvoltage 825. Thus, in this embodiment, the control circuit 820 is biasedvia the regenerative snubber 810.

Thus, a significant portion of the energy that would otherwise bedissipated as heat in a lossy snubber, e.g. resistor and capacitorsnubber, can be “recaptured” and utilized. As illustrated in FIG. 8, theenergy can be utilized to bias the control circuit 820. In other words,the snubber and the bias circuitry can be combined into a singlecircuit. In addition, depending on the design of the snubber and biasavailable from the snubber, other items in a system could be powered viathe snubber circuitry. For example, in a device dissipating a largeamount of heat which requires a cooling fan, the cooling fan, inaddition to or in lieu of the control circuit, could be powered by theregenerative snubber.

FIG. 9 illustrates a regenerative snubber, in accordance with oneembodiment. MOSFETs Q1 942 and Q2 940 and their corresponding explicitanti-parallel transistor diodes 928 918 form an AC MOSFET switch aspreviously described. When the current i 990 flows as illustrated, andQ1 942 and Q2 940 are turned off, e.g. the circuit enters a turn-offstate, the current is diverted through energy storage device C₁ 910 andcapture circuitry R₁ 912 and d₂ 914. This diversion causes charge tobuild on an energy storage device, bias capacitor C₃ 916. Bias capacitorC₃ 916 provides a bias voltage between a bias node 905 and a ground 950for the bias circuit. Current then continues through explicit transistordiode 918 of Q₂ 940. When Q1 942 and Q2 940 are turned back on, C₁ 910is reset. That is, the charge stored on C₁ 910 is discharged by flowingthrough Q1 942, d₁ 970 and is then dissipated in R₁ 912.

The symmetry of the snubber/biasing circuit allows for the charge tooccur with both directions of AC flow. When the current 990 is reversedand Q₁ 942 and Q₂ 940 are turned off, the flow is through devices C₂920, R₂ 922, d₃ 924, charging C₃ 916 and then through explicittransistor diode 928 of Q₁ 942. When Q₁ 942 and Q₂ 940 are turned backon, C₂ 920 is reset and the charge stored on capacitor C₂ 920 flowsthrough MOSFET Q₂ 940, d₄ 972 and is dissipated in R₂ 922. Thus, duringthe turn-off period of the AC MOSFET switch, charge is supplied to biascapacitor C₃ 916 resulting in bias voltages at bias node 905. Thevoltage between the ground 950 and bias node 905 provides bias for thecontrol circuit.

FIG. 10 illustrates a regenerative snubber, in accordance with anotherembodiment. Capacitor C₃ 1016, stores charge that can be utilized tobias a control circuit. When MOSFETs Q₁ 1042 and Q₂ 1040 are turned off,the current i 1090 flows through C₁ 1010 and d₂ 1014 and charges C₃1016. The current continues through explicit transistor diode 1018 of Q₂1040. In this embodiment, there is no resistor in the turn-off circuitto dissipate energy. Thus, during turn off, more energy may be deliveredto charge C₃ 1016.

When MOSFETs Q₁ 1042 and Q₂ 1040 are turned on, C₁ 1010 resets throughQ₁ 1042, d₁ 1070 and R₁ 1012. When current i flow 1090 reverses, similarresults occur through snubbing/biasing devices C₂ 1020, R₂ 1022, d₄1072, explicit transistor diode 1028 and d₃ 1024.

FIG. 11 illustrates a regenerative snubber, in accordance with yetanother embodiment. By modifying the embodiment of FIG. 10, andreplacing the resistors with inductors L₁ 1113 and L₂ 1123, the energyloss during reset can also be greatly reduced allowing significantlymore of the snubbed energy to be captured and pumped to C3 1116. When Q₁1142 and Q₂ 1140 are turned off, capacitor C3 1116 is charged througheither C₂ 1120 and d₃ 1124 or C₁ 1110 and d₂ 1114, as previouslydiscussed, depending on the current direction through the MOSFETs at thetime of the turn off. Assume current flow i 1190, when Q₁ 1142 and Q₂1140 are turned on. The charge stored on C₁ 1110 causes current to flowthrough L₁ 1113. The L₁C₁ circuit will resonate at a frequency which maybe expressed asω₀=½π√{square root over (L ₁ C ₁)}To provide adequate snubber reset, the resonant frequency of L1 C1 andL2 C2 can be chosen such that the frequency is at least as high as theminimum period expected for conduction Q1 1142 and Q2 1140.

When Q₁ 1142 and Q₂ 1140 turn on the resonance of L1 C1 results in anattempt to invert the voltage on C₁ 1110. When the voltage at the anodeto d₂ 1114 reaches a potential just above that of bias node 1105, d₂1114 switches on allowing additional energy to pump into C3 1116. Thisembodiment advantageously reduces the amount of energy loss by removingresistors from both the turn-off and reset operation of the snubber/biascircuit.

Also illustrated in FIG. 11 are snubbers for the active devices of thesnubber/bias circuitry. The circuit contains a number of diodes whichmay themselves be a source of conducted and radiated emission to thecircuit. In order to facilitate the reduction of these conducted andradiated emissions, RC snubber circuits 1180 can be placed across thediodes.

In one embodiment, fast switching diodes are utilized in thesnubber/biasing circuit. For example, diodes with switching time of 10ns or faster may be utilized in one embodiment.

When the AC MOSFET is switching, levels of bias current provided by thecircuit will be at relatively high levels compared to when the AC MOSFETis not switching. For example, assuming the AC MOSFET switch isoperating at 28.5 kHz, with a line voltage of 120 V_(RMS) and 0.01μFarad capacitance for C₁ and C₂. Each of the snubber capacitorseffectively “sees” the RMS voltage across it with C₁ 1110 seeing thefirst half cycle and C₂ 1120 seeing the second half cycle. The snubbercapacitors are charging and discharging at the switch frequency. Thecurrent available to charge C₃ can be calculated as follows:

$\begin{matrix}{Q = {{i \times t} = {c \times v\text{~~~=}\text{>}}}} \\{i = {{( {c \times v} )/t} = {c \times v \times f}}} \\{i = {( {0.01 \times 10^{6}} )(120)(28500)}} \\{i = {34.2\mspace{14mu}{mA}}}\end{matrix}$This value may be doubled in the embodiment in which an inductor is usedto invert the voltage of the snubber capacitor during snubber reset.

However, when the AC MOSFET switch is idle, the switching of the snubbercircuit occurs with the line frequency of, for example, 50-60 Hz. Inthis case, the capacitor C3 1116, which see the peak value of V, willhave much less current to charge it:i=(0.01×10⁶)(120×√{square root over (2)})(60)i=0.10 mA

FIG. 12 illustrates a regenerative snubber, in accordance with anotherembodiment. In this embodiment, two series resistors R₃ 1282 and R₄ 1284are added along with a full waver rectifier 1280. These elements may beutilized to help provide additional DC bias. This additional DC bias maybe useful, when the circuit is idle, in supplying additional charge tobias capacitor C₃ 1216. For example, as previously noted, assuming apower source of 120 VAC_(RMS), the resistors R₃ 1282 and R₄ 1284, at 60kΩ, provide an additional:(120)/(60 k)=2.0 mAThus, by placing the full wave rectifier 1280 and series resisters R₃1282 and R₄ 1284 in the circuit as illustrated, the current available tothe capacitor C₃ 1216 for providing bias to the control circuitry, whilethe AC MOSFET switch is idle, can be increased from 0.1 mA to 2.1 mA.

FIG. 13 illustrates a regenerative snubber with additional DC bias, inaccordance with another embodiment. In the circuit, in addition to thecurrent supplied by capacitors C₁ 1310 and C₂ 1320, to charge C₃ 1316,resistors R₁ 1388 and R₂ 1386 are utilized to provide increased currentto charge C₃ 1316. Similar to the calculations above, utilizing 60 kΩresistors for R₁ 1388 and R₂ 1386 results in an additional 2.0 mA ofcurrent being available. This increases the bias current to 2.1 mA.

Also illustrated in FIG. 13 is the use of a zener diode 1384 across C₃1316. It is possible that the energy stored on C₃ 1316 may cause thevoltage at the V_(BIAS) node 1305 to rise to levels that exceed what isallowed by a control circuit biased by the regenerative snubber. In thiscase, by placing a zener diode 1384 with the proper breakdown voltageacross the capacitance device C₃ 1316, a proper voltage value can bemaintained at the bias node 1305. For example, if a V_(BIAS) value for acontrol circuit of 13 volts is desired, a zener diode with a 15 voltbreakdown voltage can be placed across the capacitance device C₃ 1316 toensure that the voltage level across C₃ 1316 does not exceed 15 volts.In an alternative embodiment, a resister is placed across C₃ 1316 tofacilitate maintenance of a voltage across C₃ 1316. In anotherembodiment an avalanche diode is utilized to ensure that a propervoltage value may be maintained at the bias node 1305.

While the previous embodiments illustrate a regenerative snubber in usewith the AC MOSFET switch, the regenerative snubber may be used in otherconfigurations. FIG. 14 illustrates a regenerative snubber in use with aDC-DC converter, in accordance with one embodiment. Illustrated in FIG.14 is an electrically isolated flyback converter. Power switch 1430 isutilized to control power delivery to the load 1425. Power switch 1430is controlled by control circuit 1470. Control circuit 1470 is biased bybias node 1405 charged by regenerative snubber 1440. While anelectrically isolated flyback converter DC switching circuit isillustrated in conjunction with the regenerative snubber, other DCswitching circuit types such as boost and buck-boost converters may beutilized.

Regenerative snubber 1440 is utilized to capture energy stored in theelectrically isolated flyback converter when power switch 1430 isswitched off. When power switch 1430 turns off, current i 1490 flowsthrough C₁ 1410 and d₁ 1414 and charges C₃ 1416 and thus correspondingbias node 1405. When power switch 1430 turns on, C₁ 1410 resets throughpower switch 1430, d₂ 1419 and L₁ 1418.

During low frequency operation of the DC-DC switching circuit,sufficient current to provide adequate bias may not be provided by C₁1410. Thus, resistor R₁ 1412 is coupled across C₁ 1410 to provideadditional bias. An appropriate value of R₁ 1412 for providing adequatebias current for bias node 1405 may be application dependant.

Thus, a unique method of providing bias for a control circuit isprovided. Although specific embodiments have been illustrated anddescribed herein, it will be appreciated by those of ordinary skill inthe art that a variety of alternative and/or equivalent embodiments maybe substituted for those disclosed herein without departing from thespirit and scope of the claimed subject matter. This application isintended to cover any adaptations or variations of the preferredembodiments discussed herein. Therefore it is intended that the presentinvention be limited only by the claims and the equivalents thereof.

1. An apparatus comprising: an AC switching circuit; a control circuitcoupled to the AC switching circuit; and a biasing snubber circuitcoupled to the switching circuit and the control circuit to captureenergy from a circuit switched by the switching circuit and to provideat least a portion of the captured energy to bias the control circuit;wherein the AC switching circuit comprises: a first Field EffectTransistor (FET) having a first source, a first gate and a first drain;a second FET having a second drain, a second source coupled to the firstsource and a second gate coupled to the first gate; a first diode havinga first anode coupled to the first source and a first cathode coupled tothe first drain; and a second diode having a second anode coupled to thesecond source and a second cathode coupled to the second drain.
 2. Asnubber circuit comprising: a first energy storage device; circuitrycoupled to the first energy storage device to facilitate capturing, bythe first energy storage device, energy of a switching circuit and tofacilitate resetting of the first energy storage device; and a secondenergy storage device coupled to the first energy storage device tostore the captured energy and to provide at least a portion of thecaptured energy to a control circuit.
 3. The snubber circuit of claim 2wherein the switching circuit is a DC switching circuit.
 4. The snubbercircuit of claim 2 wherein the switching circuit is an AC switchingcircuit.
 5. The snubber circuit of claim 2 wherein the circuitrycomprises a plurality of diodes.
 6. The snubber circuit of claim 2wherein the second energy storage device provides a bias source for thecontrol circuit of the switching circuit.
 7. The snubber circuit ofclaim 2 wherein the second energy storage device provides a bias sourcefor a fan.
 8. The snubber circuit of claim 2 wherein at least one of thefirst and second energy storage devices comprises a capacitor.
 9. Thesnubber circuit of claim 2 wherein at least one of the first and secondenergy storage devices comprises an inductor.
 10. A method of supplyingpower to a control circuit comprising: capturing energy of a switchingcircuit in a first energy storage device; providing at least a portionof the captured energy in the first energy storage device to a secondenergy storage device; and providing at least a portion of energy storedon the second energy storage device to power the control circuit. 11.The method of claim 10 wherein the first circuit comprises a controlcircuit for the switching circuit.
 12. The method of claim 10 whereinthe switching circuit comprises an AC switching circuit.
 13. The methodof claim 10 wherein the switching circuit comprises a DC switchingcircuit.
 14. A snubber circuit to power a first circuit comprising:means for capturing energy of a switching circuit in a first energystorage device; means for providing at least a portion of the capturedenergy in the first energy storage device to a second energy storagedevice; and means for providing at least a portion of energy stored onthe second energy storage device to power the first circuit.
 15. Thesnubber circuit of claim 14 wherein at least one of the first energystorage device and the second energy storage device comprise capacitors.16. The snubber circuit of claim 14 wherein the first circuit comprisesa control circuit for controlling the switching circuit.
 17. Anapparatus comprising: an AC switching circuit; a control circuit coupledto the AC switching circuit; and a biasing snubber circuit coupled tothe switching circuit and the control circuit to capture energy from acircuit switched by the switching circuit and to provide at least aportion of the captured energy to bias the control circuit; wherein theAC switching circuit includes a first snubbing capacitor and a first;current limiting device, wherein the AC switching circuit is configuredto switch the first current limiting device into circuit when the firstsnubbing capacitor is reset.
 18. The apparatus of claim 17, wherein thefirst current limiting device comprises a resistor.
 19. The apparatus ofclaim 17, wherein the first current limiting device comprises aninductor.
 20. The apparatus of claim 17, wherein the AC switchingcircuit includes a second snubbing capacitor and a second currentlimiting device, wherein the AC switching circuit is configured toswitch the first current limiting device into circuit when the firstsnubbing capacitor is reset during a positive half cycle and wherein theAC switching circuit is configured to switch the second current limitingdevice into circuit when the second snubbing capacitor is reset during anegative half cycle.
 21. The apparatus of claim 17, wherein the firstcurrent limiting device comprises an inductor and wherein the ACswitching circuit is configured to pump charge during snubbing ofcurrent and during reset of the first snubbing capacitor.
 22. Anapparatus comprising: a switching circuit; a control circuit coupled tothe switching circuit; and a biasing snubber circuit coupled to theswitching circuit and the control circuit to capture energy from acircuit switched by the switching circuit and to provide at least aportion of the captured energy to bias the control circuit, wherein theswitching circuit includes a first snubbing capacitor and a firstcurrent limiting device, wherein the switching circuit is configured toswitch the first current limiting device into circuit when the firstsnubbing capacitor is reset.
 23. The apparatus of claim 22, wherein thefirst current limiting device comprises a resistor.
 24. The apparatus ofclaim 22, wherein the first current limiting device comprises aninductor.
 25. The apparatus of claim 22, wherein the switching circuitincludes a first transistor and a second transistor, the firsttransistor and the second transistor having source terminals connectedin common.
 26. The apparatus of claim 22, wherein the switching circuitcomprises an AC switching circuit including a second snubbing capacitorand a second current limiting device, wherein the switching circuit isconfigured to switch the first current limiting device into circuit whenthe first snubbing capacitor is reset during a positive half AC cycleand wherein the switching circuit is configured to switch the secondcurrent limiting device into circuit when the second snubbing capacitoris reset during a negative half AC cycle.
 27. The apparatus of claim 22,wherein the first current limiting device comprises an inductor andwherein the switching circuit is configured to pump charge duringsnubbing of current and during reset of the first snubbing capacitor.28. The apparatus of claim 22, wherein the switching circuit isconfigured to supply initial power to the control circuit.